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Design and Experimental Investigation of a Compact Circularly Polarized Integrated Filtering Antenna for Wearable Biotelemetric Devices Zhi Hao Jiang, Member, IEEE, Micah D. Gregory, and Douglas H. Werner, Fellow, IEEE

Abstract—A compact circularly polarized (CP) integrated filtering antenna is reported for wearable biotelemetric devices in the 2.4 GHz ISM band. The design is based on a mutual synthesis of a CP patch antenna connected to a bandpass filter composed of coupled stripline open-loop resonators, which provides an integrated low-profile radiating and filtering module with a compact form . The optimized filtering anfactor of , tenna is fabricated and measured, achieving an an axial ratio of less than 3 dB and gain higher than 3.5 dBi in the targeted ISM band. With the integrated filtering functionality, the antenna exhibits good out-of-band rejection over an ultra-wide frequency range of 1–6 GHz. Further full-wave simulations and experiments were carried out, verifying that the proposed filtering antenna maintains these desirable properties even when mounted in close proximity to the human body at different positions. The stable impedance performance and the simultaneous wide axial ratio and radiated power beam widths make it an ideal candidate as a wearable antenna for off-body communications. The additional integrated filtering functionality further improves utility by greatly reducing interference and crosstalk with other existing wireless systems. Index Terms—Body-area network (BAN), wearable antennas, wearable biotelemetric device, wireless off-body communication.

I. INTRODUCTION

B

ODY-CENTRIC wearable wireless communication systems hold great promise for their rich potential in diverse areas (e.g., medical monitoring, firefighter tracking, wearable computing, battlefield survival, etc.) eliciting considerable research interest and industrial investment over the past decade [1], [2]. Among these applications, biotelemetry is the one that may have the most widespread and profound impact on the daily life of all human beings [3]. Such a system involves a remote central processing terminal and a group of sensor nodes placed at different positions inside/on the human body, which monitor various bio-signals of a person such as blood pressure, heart

Manuscript received January 30, 2015; revised May 20, 2015; accepted May 23, 2015. This work was supported by the National Science Foundation ASSIST Nanosystems ERC under Award Number EEC-1160483. This paper was recommended by Associate Editor D. Cumming. The authors are with the Electrical Engineering Department, The Pennsylvania State University, University Park, PA 16802 USA (e-mail: zuj101@psu. edu; [email protected]; [email protected]). Color versions of one or more of the figures in this paper are available online at http://ieeexplore.ieee.org. Digital Object Identifier 10.1109/TBCAS.2015.2438551

rate, glucose level, electrical activity of the brain and heart, etc. At the heart of the system is a battery-powered compact wearable unit, which is responsible for the collection and temporary storage of information measured by implanted/on-body sensors and the wireless data transmission to the remote central terminal where doctors can access the data. While much advancement has been made on the miniaturization and sensitivity improvement of wearable/implanted sensors [4], it is the wearable processing/computing unit that plays the most critical role in the system since it provides the vital link between the patients and doctors. As the complexity and the amount of biomedical information to be acquired from a body-area network biotelemetry system become more demanding, the wearable unit therefore needs to function in a low power mode in order to maintain a sufficiently long operation time [3]. In such a circumstance, considerable efforts have been carried out to enable a low-power but reliable wireless link between the wearable device and a remote central terminal [5], [6]. As a critical component in the system, the wearable antenna for offbody communications provides the physical mechanism that enables transmission and reception of wireless signals in a bodyarea network. Therefore, high-performance, unobtrusive wearable antennas are critically important to the successful creation of a reliable and power-efficient link for the wireless exchange of data. At the same time, they can also mitigate the stringent power requirements imposed on other components in the wearable transceiver since the total system gain to enable proper functioning would remain the same [1], [7]. Because the human body is naturally in close proximity to these devices, the loading effect caused by lossy tissue makes the design of a highly efficient wearable antenna extremely challenging. Simultaneously, it is also desirable for these antennas to be light-weight and low-profile, as well as having minimal electromagnetic impact to the human body which is characterized by the specific absorption rate (SAR) [1], [8]. So far, numerous wearable antenna configurations have been proposed for off-body communications in the 2.4–2.48 GHz industrial, scientific and medical (ISM) band. The earliest examples, such as vertical monopole [9] and inverted-F antenna designs [10], are not low profile or conformal. Planar microstrip monopoles [7], [11], [12] and planar inverted-F antennas [13], [14], including single-band, multi-band, and wideband versions that cover the ISM band, have small form factors but undesirably radiate a significant amount of power into the human body, resulting in low efficiency. Hence, they

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are usually positioned a certain distance away from the human body. Patch antennas [15] and cavity-backed slot antennas [16], [17] are amenable for off-body communications due to their broadside radiation patterns, but they exhibit a narrow bandwidth and ground plane size dependent front-to-back (FB) ratios, as well as an often observed tilted main beam. Isotropic artificial magnetic conducting (AMC) surfaces [18], [19] and a recently proposed strongly-truncated anisotropic metasurface [20] have also been employed to obtain a high degree of isolation between the antenna and human tissue while maintaining a reasonably small overall profile. However, almost all of these wearable antennas are linearly polarized, which can potentially lead to unreliable wireless links due to constant human body motion in realistic environments. Circularly polarized (CP) antennas, on the other hand, tend to be more favorable for wearable applications due to their improved signal robustness to human body movement and multipath interference [21], [22]. As mentioned before, in many power-efficient wearable biotelemetric devices, sensitive receivers along with low-power circuits are often utilized which allow for a longer battery life [23], [24]. Due to the ubiquitous wireless signals distributed in the 1–6 GHz spectrum, interference and crosstalk may deteriorate the performance of these wearable receivers and can even potentially damage or destroy their electronic components. Therefore, a bandpass filter (BPF) is necessary at the front-end to discriminate the desired signal from extraneous signals outside the targeted band [25], [26]. In conventional approaches, the BPF and the antenna are designed separately and then directly cascaded by sharing a common reference impedance or are both connected to a two-port matching circuit, typically resulting in degraded performance and increased device footprint. A mutual synthesis method for designing the BPF and antenna as a single module provides an alternative and more attractive solution [27]. So far, several filtering antennas have been investigated by integrating various filter structures with different types of antennas [28]–[37]. These examples demonstrate that co-designed filtering antennas indeed have smaller overall device volume and superior band selectivity. These proposed filtering antenna structures are not suitable candidates for wearable applications, however, due to their large footprints, high profile, and linear polarization. In addition, most of these demonstrated planar filtering antennas possess near-isotropic radiation patterns and/or require fabrication of complicated 3D structures. In this paper, we propose and experimentally demonstrate a CP wearable antenna integrated with microwave BPF circuits, providing a compact module for body-worn devices in the ISM band. In Section II, we describe the co-design methodology for the filtering antenna by illustrating the layout, equivalent circuit, structural design, and optimized results. Section III presents a comparison among the proposed filtering antenna, the antenna without the filter, and an independently designed and connected antenna with a BPF. Section IV gives the measured results in free space. Numerical and experimental on-body performance investigations are provided in Section V, followed by the conclusions in Section VI.

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Fig. 1. (a) Exploded view of the proposed CP filtering antenna using coupled open-loop resonators with stripling feeding. (b) Stacked side view of the filtering , , , , antenna. The dimensions are , , , all in millimeters. All the substrates are and . The copper layers Rogers RT/duroid 5880 with thick. are 18

II. WEARABLE FILTERING ANTENNA SYNTHESIS A. Topology of the CP Filtering Antenna The configuration of the wearable CP filtering antenna is illustrated in Fig. 1, which consists of a nearly square patch radiator with truncated corners on the top and a planar stripline microwave BPF circuit on the bottom. The filter and the patch are connected through a metallic pin that passes through the top ground plane of the stripline circuit. The feeding position of the pin on the patch is offset from the center in order to obtain low CP axial ratio (AR). The top ground plane of the filter also serves as the bottom ground plane of the patch radiator. The BPF circuit consists of four edge-coupled stripline open-loop resonators (SOLRs) [38], where the signal passes sequentially from resonator 1 to resonator 4. The filtering antenna is fed by a subminiature version A (SMA) connector from the side, which enables the antenna to be placed very close to the surface of

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Fig. 2. Equivalent circuit model of the CP filtering antenna.

human bodies compared to the bottom-fed topologies considered in previous CP wearable patch antennas [21], [22]. The form factor of the filtering antenna is , which is smaller than many previously demonstrated ISM band wearable antennas [15]–[22]. B. Equivalent Circuit Model The equivalent circuit of the integrated filtering antenna is shown in Fig. 2. The CP patch radiator is modeled as two series-connected parallel RLC resonators ( and ) to account for the orthogonal modes in the and directions, respectively. A series inductor is used to model the feeding pin while the shunt is employed to represent the parasitic capacitor of the circular stripline open-end. The pin-fed patch and the BPF are connected by a transmission line (TL) with a propagation constant , a characteristic impedance , and a length . Within the narrow ISM band, the input impedance of the TL connected pin-fed patch behaves like a simple load with dispersive complex impedance. The four parallel RLC resonators represent the four SOLRs, while the direct and cross coupling between them are modeled as admittance inverters ( , ) [39]. Weak cross coupling between resonators 1 and 3 as well as 2 and 4 exist due to the compact layout of the BPF structure. These coupled resonators can be either synchronously or asynchronously tuned. The input impedance of the BPF filter, which is also the input impedance of the entire filtering antenna, is set to match a 50 SMA connector. Conventionally, the BPF and the antenna are designed separately by matching both of them to a common reference impedance such as 50 or 75 . An additional impedance matching circuit is usually required in order to obtain the desired performance, which significantly increases the complexity of the system in terms of weight, size and losses. Here, the output impedance of the BPF and the input impedance of the TL connected antenna component are designed to be complex conjugates [27]. During the mutual synthesis process, the TL loaded antenna and the filter portions are first treated separately but designed simultaneously in order to obtain conjugate matching and good radiating properties in the desired band. They are then connected and optimized to achieve the best performance for the overall system. C. Radiating Patch and Stripline Filter Design The detailed geometry of the pin-fed radiating patch is depicted in Fig. 3(a). In order to support circular polarization over a wide angular range, the pin is offset from the center of the

Fig. 3. (a) Top view of the patch radiator layer. The optimized dimensions are , , , , , , , all in millimeters. (b) Input impedance response of the pin fed CP patch obtained from the full-wave simulation and equivalent , circuit model. The corresponding lumped circuit element values are , , , , , , , .

patch and the corners are slightly truncated. The Ansoft high frequency structure simulator (HFSS) predicted input impedance of the pin-fed CP patch radiator is shown in Fig. 3(b), where two resonant modes at 2.375 and 2.525 GHz are observed, corresponding to the resonating modes in the and directions, respectively. The impedance calculated using the equivalent circuit model agrees well with the full-wave result in the 2.1–2.8 GHz frequency range except for a narrow region around 2.72 GHz due to the excitation of a weak parallel plate waveguide mode. The topology of the coupled-resonator BPF section of Fig. 2 is illustrated in Fig. 4(a). The resulting scattering parameters of the BPF alone can be readily calculated using the coupling matrix

(1) and the external quality factors of the first and last resonators and . A fourth order Chebyshev BPF with a 0.1 dB in-band ripple and a fractional bandwidth (FBW) of 4.09% centered at 2.445 GHz was employed as the base design, which has , , , , where all the diagonal elements are zero [26]. In order to obtain a passband in the ISM band with sharp roll-offs when the pin-fed patch and BPF are connected by a TL as an

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Fig. 4. (a) Coupling topology of the coupled resonator BPF. (b) Top view of , , the stripline BPF layer. The optimized dimensions are , , , , , , , , , , , , , , , , , , , , , , , all in millimeters.

integrated device, the coupling coefficients in and the external quality factors based on the Chebyshev design, as well as the length of the connecting TL were optimized using the covariance matrix adaptation evolutionary strategy (CMA-ES) [40], [41]. This real-valued optimization algorithm is well suited to the design optimization problem considered here. Its operation proceeds by reshaping and moving a multi-variate Normal search distribution within a multi-dimensional parameter space. The search distribution automatically aligns itself to the transverse along the contour of decreasing cost, thereby expanding or shrinking its extent during each iteration in order to find the optimal solution with a reduced total number of function calculations. Therefore, it has been proven to be more efficient than many previous widely used global optimization methods such as genetic algorithm and particle swarm optimization [41]. Here, a conjugate matching is desired between and , where can be readily calculated from the TL equation and the previously obtained [39]. The optimized external quality factors are and , while the coupling matrix becomes

(2)

indicating that the four resonators are asynchronously tuned. In addition, a value of 30.5 mm has been chosen for . and have opposite signs due to the fact that one is primarily magnetic coupling while the other is electric. The physical stripline BPF implemented using the coupled resonators can be determined using the procedures given in [26] by simulating a single and a pair of SOLRs, which can be used as the values of an initial design. It should be noted that the equivalent circuit shown in Fig. 2 is valid only around the fundamental resonant frequencies of the coupled resonators. The possible spurious passband at the harmonic frequency of the resonators, i.e.,

Fig. 5. (a) The obtained from full-wave simulation, lumped circuit model, and free-space measurement of the prototype. (b) Simulated and measured gain in free space. (c) Simulated and measured broadside axial ratio in free space.

4.8 GHz, is suppressed due to the fact that the physical admittance inverters implemented by the electric and magnetic coupling between the SOLRs no longer have the required values to form a passband. Thus, in the entire range from 1 to 6 GHz, only a single passband is available, which will be presented in Section III. D. Optimized Filtering Antenna Results When all the components are integrated, mutual coupling between each adjacent resonator pair may cause minor frequency shifts and/or coupling parameter variations. Again, a MATLAB based CMA-ES optimization code was employed and coupled with HFSS to fine tune the geometrical dimensions for the integrated filtering antenna module in order to achieve performance resembling that obtained from the circuit model. The of the optimized design is shown in Fig. 5(a), where the full-wave simulation and the circuit model are in good agreement with a maximum of 13.7 dB and a 3.9% fractional bandwidth in the passband centered at 2.44 GHz. Its broadside gain [see Fig. 5(b)] has a profile similar to the transmission of the BPF with a peak gain of 4.7 dBi and in-band gain variation of 1.1 dB. The averaged radiation efficiency in the passband is around 58%, indicating that the insertion loss of the added BPF causes a drop of 32% in the radiation efficiency. Its gain is at least 20 dB lower than the peak in the frequency regimes below 2.3 GHz and above 2.56 GHz, indicating roll-offs of 237.5 dB/GHz and 271.4 dB/GHz, respectively. The broadside AR displayed in Fig. 5(c) is below 3 dB from 2.409 to 2.467 GHz, which is nearly identical to that of the pin fed patch alone. The filtering

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Fig. 6. Simulated and measured free-space normalized total power radiation patterns at 2.445 GHz in (a) the x-z plane and (b) the y-z plane. Fig. 8. Simulated (a) and (b) gain of the integrated filtering antenna and the antenna without the BPF. The highlighted regions denote some major wireless bands for existing systems.

Fig. 7. Simulated (a) reflection magnitude and (b) reflection phase at the BPF and antenna connecting ports.

antenna has a maximum total radiated power at broadside in both the and planes with nearly equal half power beam widths (HPBWs) of 86 and 87 , respectively, as shown in Fig. 6. A front-to-back ratio of about 17.5 dB is achieved. The reflection coefficients at the input port of the TL connected pin-fed patch and the output port of the BPF are compared in Fig. 7. One can observe that none have well-matched in-band impedance. However, due to the fact that they are nearly conjugated, i.e., and ∠ –∠ , the integrated filtering antenna achieves a much better return loss in the targeted band, which corroborates previous findings for bulky cavity-based filtering antenna designs in the Ku-band [27]. III. COMPARISON WITH REGULAR CIRCULARLY POLARIZED PATCH ANTENNA W/WO DIRECTLY CASCADED BPF To demonstrate the advantages of the mutual synthesis method in designing filtering antennas, the performance of the proposed integrated antenna was compared to a CP patch antenna without a filter, as well as to a CP patch directly connected to a separately designed BPF using the conventional method. As presented in Fig. 8, the performance of the CP patch antenna with and without the BPF are compared within an ultra-wide frequency range of 1 to 6 GHz. The integrated filtering antenna has a single passband within the targeted frequency range and stopbands elsewhere, except for a narrow

shallow resonance around 4.8 GHz, due to the harmonic resonance of the BPF. However, the drops only down to 4 dB because the coupling parameters between resonators no longer have the values required to form a passband at this high frequency band, which causes the impedances of the TL-loaded pin-fed patch and the BPF to lose their conjugate match. Its gain (not including the impedance mismatch) also has a single narrow peak at the targeted band, with an out-of-band rejection of more than 25 dB. In contrast, the patch antenna alone has a wide band around the targeted range from 2.4 to 2.6 GHz with larger reflection than that of the filtering antenna within the ISM band. Other narrow and wide spurious bands can also be observed at around 1.8 GHz, 3.7 GHz, and 4.5–5.9 GHz. Its gain also has a profile much flatter than that of the filtering antenna, well above 10 dBi within all of the spurious bands, and even higher than 0 dBi in some places. The poor selectivity of both the and gain makes the conventional CP patch antenna subject to interference and crosstalk caused by other existing wireless systems such as the various GPS bands between 1 and 2 GHz, the 1700–1900 MHz GSM band, the 1700–2100 MHz UMTS band, the 2100 and 2600 MHz LTE bands, the 3.4–3.6 WiMax band, the 3.6–3.7 GHz and 4.9–5.8 GHz WLAN bands, the 4.2–4.4 GHz aeronautical radio band, and so on, as highlighted in Fig. 9. The realized gain values of the filtering antennas in the stopbands and passband will have even greater contrast when both the impedance mismatch and gain are taken into account. This comparison shows that the filtering antenna achieves a much better spectrum selectivity than the patch antenna alone, consequently yielding greatly reduced interference and crosstalk which is especially vital for low-power sensitive wearable receivers. Particularly important is the fact that even with all of the aforementioned performance enhancements, the device form factor remains the same. The second study compares the proposed filtering antenna to a CP patch antenna directly connected to a BPF which are both designed separately to match to 50 . The designed BPF alone achieves an in-band of less than 15 dB. As displayed in Fig. 10, the directly cascaded patch and BPF has a slightly broader passband along with a much higher than that of the integrated filtering antenna obtained via the mutual synthesis method. More specifically, in the frequency range from 2.45

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TABLE I PROPERTY COMPARISON AMONG DIFFERENT 2.4 GHZ ISM BAND WEARABLE ANTENNA DESIGNS

bandwidth in percentage.

IV. FREE-SPACE MEASUREMENT RESULTS

Fig. 9. Simulated of the integrated filtering antenna and the antenna directly connected with the BPF. In the latter case, the antenna and the BPF are designed separately to achieve the best matching to 50 .

Fig. 10. Photographs of (a) the stripline BPF filter circuit and (b) the assembled filtering antenna module.

to 2.49 GHz, the of the directly cascaded patch and BPF is even above 10 dB. This comparison illustrates the importance of the mutual synthesis method employed when designing a BPF and antenna as a single integrated module.

The components of the integrated filtering antenna were fabricated by a standard printed circuit board etching method and then assembled to form the multilayer device. Nine nylon screws were used on the outer periphery of the filtering antenna in order to achieve a tight bonding of layers. A SMA connector was soldered to the stripline feed and corresponding ground layers. Fig. 10 shows photographs of the fabricated stripline BPF structure and the final assembled module. The impedance of the filtering antenna was measured using an Agilent E8364B network analyzer. As shown in Fig. 5(a), the measured exhibits good filtering properties with from 2.385 to 2.485 GHz. Good agreement is achieved in terms of the position of reflection zeros, out-of-band rejection levels, and passband width. The minor difference in the in-band and slightly decreased roll-offs can be attributed to fabrication and assembly inaccuracy. The measured broadside gain is plotted in Fig. 5(b), which has a flat passband with values higher than 3.5 dBi in the targeted band and high rejection elsewhere. The gain in the passband is about 1.1 dB lower than the simulated value primarily due to the loss caused by the relatively low conductivity solder near the SMA and the vertical pin. The measured broadside AR is below 3 dB from 2.417 to 2.473 GHz, which has a slight shift of about 5 MHz from the simulated value. The normalized far-field patterns of the total radiated power in both the x-z and y-z planes (see Fig. 6) measured in an anechoic chamber correspond well to simulated predictions with only a slightly larger backlobe (about 1.5 dB higher). Above all, the measurements confirmed a fairly high-fidelity implementation of the proposed filtering antenna design. In order to compare with the previously reported 2.4 GHz ISM band wearable antennas, the antenna properties including footprint, antenna profile, bandwidth, polarization, and pattern type are listed in Table I.

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Fig. 11. Configuration of the filtering antenna mounted on (a) the arm, (b) the chest, and (c) the back of the human body model in HFSS. Simulated and measured and gain, (g)–(i) AR, and (j)–(l) total power radiation patterns in both the horizontal x-y and vertical (x-z or y-z) planes of the integrated filtering antenna (d)–(f) mounted on the arm, chest, and back of a full-scale human body model.

V. NUMERICAL AND EXPERIMENTAL INVESTIGATION ON-BODY PERFORMANCE

OF

In order to evaluate the performance of the proposed integrated filtering antenna when it is mounted directly on a human body as a wearable antenna for off-body communications, both full-wave numerical and experimental investigations were carried out, the results of which are presented in this section. A. Impact of the Human Body on Antenna Performance First, the impact of the human body on the input impedance and the radiation properties of the filtering antenna was studied. Here, a homogenous full-scale human body model with a height of 174 cm and a chest width of 46 cm was adopted [42]. The HFSS integral equation (IE) solver based on the method of moments was employed where the human body model was assigned as the IE region and the filtering antenna as the default finite element region. The permittivity of the homogeneous human body model was set to be two-thirds of the permittivity of muscle, as is common practice in the literature [1], [43]. Such a homogeneous model allows for a numerically economical way to provide reasonably accurate evaluation of the impedance and radiation performance of wearable antennas [12]. The proposed filtering antenna was placed at three different locations including the arm, the chest, and the back with an antenna to body gap of approximately 2 mm, as shown in Fig. 11(a)–(c). It should be noted that the demonstrated antenna was built on a semi-rigid substrate, making it only wearable on

parts of the human body that are locally flat. A flexible wearable filtering antenna can be fabricated by printing the copper or conductive ink on polyimide substrate with high precision. Fig. 11(d)–(f) displays the and gain of the filtering antenna when placed at different positions on the body. Owing to the employed stripline BPF structure, the close proximity of the human body does not have a major impact on the electromagnetic properties of both the patch antenna and the BPF circuit, thus the remained below 12.5 dB in the targeted band. Only a slight variation in the can be observed between the filtering antenna in free-space and on-body. The gain profiles for all three cases are also well maintained with a peak value of 4.65, 4.69, and 4.68 dBi for the on-arm, -chest, and -back cases, respectively, with in-band variations all smaller than 1.2 dB. The broadside AR is displayed in Fig. 11(g)–(i) as a function of frequency, showing that a 60 MHz band remains within the targeted range. Due to the shadowing effect, a band shift of around 5 MHz occurs in the AR when the filtering antenna is located at different positions on the human body. In addition to the broadside AR, the AR curves at the center frequency of the ISM band, i.e., 2.44 GHz, are also presented in Fig. 11(g)–(i) as a function of the angular deviation from broadside in both the horizontal (x-y) and vertical (x-z or y-z) planes. Relatively wide angular ranges within which the AR is below 3 dB are achieved for all three cases. The horizontal/vertical plane beam widths for the three antenna positions are 102 /161 , 104 /141 , and 127 /131 , respectively. The

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normalized horizontal and vertical plane total power radiation patterns at 2.445 GHz for all three different positions are shown plotted in Fig. 11(j)–(l). It can be seen that the majority of the energy is directed into the hemisphere away from the human body. Slight differences exist between the radiation patterns for the antenna in free space versus when mounted on-body. This includes pattern asymmetry and radiation nulls in the backward directions which are primarily attributed to the human body shadowing effect. The horizontal/vertical plane HPBWs for the filtering antenna at the three positions are 87 /86 , 89 /88 , and 93 /94 , respectively. Importantly, the simultaneous wide angular power and AR coverage indicates that wireless links can be obtained for off-body communications which are more robust and tolerant to human body movement. Corresponding measurements were carried out where the fabricated filtering antenna was mounted on the arm, chest, and back of a person. The measured curves agree well with the simulation predictions, showing a lower than 12.5 dB within the targeted passband and slightly decreased quality factor [see Fig. 11(d)–(f)]. The gain and AR at broadside were also measured in an anechoic chamber. The gain curves, shown in Fig. 11(d)–(f), possess similar profiles compared to those of the filtering antenna in free space, though with a peak that is around 1–1.5 dB lower than the simulated results. The broadside AR curves all show a minimum at 2.443 GHz, with an bandwidth of around 60 MHz lying within the targeted ISM band [see Fig. 11(g)–(i)]. These experiments verify that the proposed filtering antenna maintains its performance when it is mounted on different positions of an actual human body. Compared to commonly used backside microstrip feeding techniques, the stripline configuration used here provides better shielding for the high-Q coupled resonator circuits. B. Surface Wave Distributions on Human Body As a byproduct, the wearable antenna also excites surface waves propagating on the human body. As shown in Fig. 12, the front and back views of the electric field distributions on the surface of the human body are displayed for the three cases shown in Fig. 11. It can be seen that, when the wearable antenna is mounted on the shoulder, the strongest surface wave is propagating along the arm. This surface wave can be utilized for on-body communication between wristband sensors, which record the heart rate, body temperature, and blood pressure, and a wearable computing terminal on the shoulder, which provides data processing and sends the information to a cloud database. When the wearable antennas are mounted on the chest or the back, the front or the back part of the torso as well as both arms are covered by surface waves, respectively. Wearable antennas at these positions can provide an on-body link with sensors on the torso which collect information such as EEG, glucose levels, and so on. In contrast to off-body communications which require a strong radiated power in order to enable wireless communication to devices that are as far as 10–20 meters, the on-body communication modes can be excited with much smaller power levels due to the fact that the distance between the wearable antennas and the on-body sensors is typically less than a half meter and the data transfer rate of each on-body sensor is

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Fig. 12. (a) Front and (b) back views of the snapshots of the electric field distribution on the surface of the human body with the proposed filtering antenna mounted on the arm, chest, and back.

low. Hence, the relatively weak surface wave, which is about 10 dB lower in power than the broadside radiation of the wearable antenna, is strong enough to support on-body communications. C. SAR Evaluation Having evaluated the impact of the human body on the performance of the filtering antenna and the surface waves excited on the human body when the antenna is placed in close proximity, we now investigate the effect of the antenna on the human body by calculating the specific absorption rate (SAR). A numerical 3D volumetric anatomical HUGO human body model produced by the National Library of Medicine was employed, which contains 40 distinct types of tissues at a mesh size resolution of [44]. The HUGO human body model was incorporated into the CST microwave studio (MWS) software package for performing SAR calculations. Similar to Fig. 11, the filtering antenna was placed at three different positions on the HUGO human body, including the arm, the chest, and the back, respectively, as shown in Fig. 13(a)–(c). As a benchmark, 100 mW of power accepted by the antenna was chosen to evaluate the SAR performance. To reduce the simulation time, only a portion of the arm, chest, and back that has a sectional area of larger than 10 times of the antenna size was used since the SAR is a near-field effect. Fig. 13(d)–(f) show the respective simulated 1g averaged SAR values for the three cases. It can be observed that, for all three cases, the peak 1g averaged SAR value is less than 0.54 W/kg, well below the 1.6 W/kg specification provided by the Federal Communication Commission (FCC) [45] and also smaller than several previously proposed compact wearable antennas [18]–[20].

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Fig. 14. Received power for a V-pol/H-pol receiving antenna with in LOS and NLOS scenario when the wearable antenna is mounted on . the chest with

Fig. 13. Configuration of the filtering antenna mounted on (a) the arm, (b) the chest, and (c) the back of the HUGO human body model in CST MWS. Simulated 1g averaged SAR values for the filtering antenna on (d) the arm, (e) the chest, and (f) the back.

D. Link Budget To evaluate the efficacy of the wearable antenna from a system point of view, we further analyzed the link budget. Considering the case of up-link off-body communication, where the wearable antenna is transmitting a signal while an antenna located at a distance away from the human body in the horizontal plane is receiving the signal. The received power can be calculated using the Friis transmission formula as (3) is the transmitted power, and are the rewhere alized gain of the transmitting and receiving antenna, respectively. is the path loss as a function of , which can in general be expressed as

(4) is the shadowing factor for the Gaussian distribution with a standard deviation and is the reference distance chosen to be 1 m [46], [47]. The is the path loss exponent. For free space environment, is 2. In a multipath propagation environment, becomes 1.5 for line-of-sight (LOS) path and 3 for non-line-of-sight (NLOS) path, respectively. We assumed the wearable antenna is mounted on the chest with a transmitted power of 10 dBm, and the receiver antenna is a well-matched linear-polarized dipole antenna with a typical gain of 2 dBi.

Fig. 14(a), (b) shows the received power as a function of d for the receiver with vertical (V, i.e., in the -direction) and horizontal (H, i.e., in the -direction) polarization at 2.41, 2.44, and 2.47 GHz, respectively. For the calculation of the link in a multipath environment, the mean antenna realized gain is used due to the fact that the system is less affected by the nulls of the radiation pattern in LOS and NLOS scenarios. It can be seen that the LOS link is much better than the NLOS link, due to the constructive reflections of the signal in the LOS environment. For the LOS case, the received power is higher than 60 dBm even at a distance of 40 meters, which well satisfies the requirement on the minimum sensitivity of standard receivers at the ISM band [48]. For the NLOS scenario, the received power is higher than 65 dBm at a distance of around 15 meters away from the human body, which can satisfy most indoor applications. Similar power level received by an H-polarized and a V-polarized antenna is attributed to the circular polarization of the wearable antenna, which makes the link robust to human body motion. At the center frequency of the band, the difference between power received by the H-polarized and V-polarized antennas are minimal, since the smallest AR occurs at such a frequency. For the down-link case, where the wearable antenna is receiving signal and the remote antenna away from the human body is transmitting, higher is allowed, which will ensure a better link range and reliability. It should be noted that, under this circumstance, the realized gain of the wearable antenna outside the targeted ISM band is extremely low due to the integrated filtering functionality, e.g., 58.9 dB at 1.5 GHz and 56.7 dB at 4.0 GHz, thereby greatly suppressing possible interference from other co-site wireless systems. VI. CONCLUSIONS In summary, we have proposed a compact microwave circuit integrated CP filtering antenna for wearable biotelemetric devices in the ISM band, with a small form factor of only . The antenna and BPF circuit are co-designed based on a mutual synthesis method rather than considered as independently designed components that are matched to a common impedance. The integrated filtering antenna module is able to effectively reject wireless signals outside the desired ISM band in the 1–6 GHz spectrum range. The antenna was fabricated and tested, exhibiting an , an axial ratio smaller than 3 dB, and a gain higher than 3.5 dBi in the targeted ISM band. The filtering antenna maintains its performance when placed at different locations on the human body, which has been verified through

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both numerical simulations and experimental investigations. The surface waves generated on the human body make possible simultaneous on- and off-body communications. The SAR evaluations based on a HUGO human body model further demonstrate that it has a very small impact on body tissue, satisfying FCC regulations. The robust input impedance and the simultaneous wide axial ratio and radiated power beam widths make the proposed filtering antenna an excellent candidate for wearable biotelemetry systems, which provides a reliable wireless link over a large range with greatly reduced interference and crosstalk between other existing wireless systems. REFERENCES [1] P. S. Hall and Y. Hao, Antennas and Propagation for Body-Centric Wireless Cummunications. Norwood, MA, USA: Artech House, 2012. [2] G. Z. Yang, Body Sensor Networks. Berlin, Germany: Springer, 2006. [3] C. S. Nikita, Handbook of Biomedical Telemetry. Hoboken, NJ, USA: Wiley-IEEE Press, 2014. [4] E. Sazonov and M. Neuman, Wearable Sensors: Fundamentals, Implementation and Applications. Waltham, MA, USA: Academic, 2014. [5] M. Contaldo, B. Banerjee, D. Ruffieux, J. Chabloz, E. L. Roux, and C. C. Enz, “A 2.4-GHz BAW-based transceiver for wireless body area networks,” IEEE Trans. Biomed. Circuits Syst., vol. 4, no. 6, pp. 391–399, Dec. 2010. [6] J. Tan, W.-S. Liew, C.-H. Heng, and Y. Lian, “A 2.4 GHz ULP reconfigurable asymmetric transceiver for single-chip wireless neural recording IC,” IEEE Trans. Biomed. Circuits Syst., vol. 8, no. 4, pp. 497–509, Aug. 2014. [7] A. Alomainy, Y. Hao, and F. Pasveer, “Numerical and experimental evaluation of a compact sensor antenna for healthcare devices,” IEEE Trans. Biomed. Circuits Syst., vol. 1, no. 4, pp. 242–249, Apr. 2007. [8] E. Moradi, K. Koski, T. Björninen, L. Sydänheimo, J. M. Rabaey, J. M. Carmena, Y. Rahmat-Samii, and L. Ukkonen, “Miniature implantable and wearable on-body antennas: Towards the new era of wireless body-centric systems,” IEEE Antennas Propag. Mag., vol. 56, no. 1, pp. 271–291, Feb. 2014. [9] P. S. Hall et al., “Antennas and propagation for on-body communication systems,” IEEE Antennas Propag. Mag., vol. 49, no. 3, pp. 41–58, Jun. 2007. [10] W. El Hajj, C. Person, and J. Wiart, “A novel investigation of a broadband integrated inverted-F antenna design; application for wearable antenna,” IEEE Trans. Antennas Propag., vol. 62, no. 7, pp. 3843–3846, Jul. 2014. [11] M. N. Suma, P. C. Bybi, and P. Mohanan, “A wideband printed monopole antenna for 2.45 GHz WLAN applications,” Microw. Opt. Technol. Lett., vol. 48, no. 5, pp. 871–873, May 2006. [12] Z. Wang, L. Z. Lee, D. Psychoudakis, and J. L. Volakis, “Embroidered multiband body-worn antenna for GSM/PCS/WLAN communications,” IEEE Trans. Antennas Propag., vol. 62, no. 6, pp. 3321–3329, Jun. 2014. [13] P. J. Soh, G. A. E. Vandenbosch, S. L. Ooi, and N. H. M. Rais, “Design of a broadband all-textile slotted PIFA,” IEEE Trans. Antennas Propag., vol. 60, no. 1, pp. 379–384, Jan. 2012. [14] Q. Bai and R. Langley, “Crumpling of PIFA textile antenna,” IEEE Trans. Antennas Propag., vol. 60, no. 1, pp. 63–70, Jan. 2012. [15] A. Alomainy, Y. Hao, A. Owadally, C. G. Parnini, Y. Nechayev, C. C. Constantinou, and P. S. Hall, “Statistical analysis and performance evaluation for on-body radio propagation with microstrip patch antennas,” IEEE Trans. Antennas Propag., vol. 55, no. 1, pp. 245–248, Jan. 2007. [16] N. Haga, K. Saito, M. Takahashi, and K. Ito, “Characteristics of cavity slot antenna for body-area networks,” IEEE Trans. Antennas Propag., vol. 57, no. 4, pp. 837–843, Apr. 2009. [17] S. Agneessens and H. Rogier, “Compact half diamond dual-band textile HMSIW on-body antenna,” IEEE Trans. Antennas Propag., vol. 62, no. 5, pp. 2374–2381, May 2014. [18] S. Zhu and R. Langley, “Dual-band wearable textile antenna on an EBG substrate,” IEEE Trans. Antennas Propag., vol. 57, no. 4, pp. 926–935, Apr. 2009. [19] H. R. Raad, A. I. Abbosh, H. M. Al-Rizzo, and D. G. Rucker, “Flexible and compact AMC based antenna for telemedicine applications,” IEEE Trans. Antennas Propag., vol. 61, no. 2, pp. 524–531, Feb. 2013.

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Zhi Hao Jiang (S’07–M’13) was born in Nanjing, China, in 1986. He received the B.S. degree in radio engineering from Southeast University, Nanjing, China, and the Ph. D. degree from The Pennsylvania State University (PSU), University Park, PA, USA, in 2008 and 2013, respectively. Currently, he is a Postdoctoral Fellow in the Computational Electromagnetics and Antennas Research Lab (CEARL) in the Department of Electrical Engineering at PSU. He was a Research Assistant in the State Key Laboratory of Millimeter Waves, School of Information Science and Engineering, Southeast University. He was with Base Station Antenna R&D, Andrew Telecommunication (China), as an Intern during the summer of 2007. His research interests include antennas, microwave circuits, metamaterials, and nanophotonics. He has coauthored three book chapters and more than 60 papers in peer-reviewed international journals and conference proceedings. Dr. Jiang was the Meritorious Winner of the 2006 Interdisciplinary Contest in Modeling, funded by the National Security Agency and administrated by the Consortium for Mathematics and its Applications. He was the recipient of the 2007 Microsoft Young Fellow awarded by Microsoft Research Asia (MSRA) and was one of the 2007 Top Ten Outstanding Students of Jiangsu Province. He won the 2012 A. J. Ferraro Outstanding Doctoral Research Award in Electromagnetics and was Honorable Mention in the 2013 IEEE AP-S International Symposium on Antennas and Propagation Student Paper Contest. He serves as a reviewer for Nature Materials, Nature Communications, Scientific Reports, IEEE TRANSACTIONS ON ANTENNAS AND PROPAGATION, IEEE ANTENNAS AND WIRELESS PROPAGATION LETTERS, IEEE MICROWAVE AND WIRELESS COMPONENT LETTERS, IEEE ANTENNAS AND PROPAGATION MAGAZINE, Nanoscale, Applied Physics Letters, Journal of Applied Physics, and PIER.

Micah D. Gregory received the B.S. degree in electrical engineering from Bucknell University, Lewisburg, PA, USA, in 2006, and the M.S. and Ph.D. degrees in electrical engineering from The Pennsylvania State University (PSU), University Park, PA, USA, in 2009 and 2013, respectively. Currently, he is working as a Research Associate and is Associate Director of the Computational Electromagnetics and Antennas Research Lab (CEARL) at PSU. His research interests include ultra-wideband and phased array antenna design, evolutionary strategies, frequency selective surfaces, and reconfigurable antennas and computational electromagnetics solvers. Other interests include parallel and high performance computer programming.

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Dr. Gregory was the recipient of the 2005 MTT-S Undergraduate/Pregraduate Scholarship Award, the 2009 A. J. Ferraro Award for Research Excellence in the Field of Antenna Engineering, and the 2013 Dr. Nirmal K. Bose Dissertation Excellence Award.

Douglas H. Werner (F’05) received the B.S., M.S., and Ph.D. degrees in electrical engineering and the M.A. degree in mathematics from The Pennsylvania State University (PSU), University Park, PA, USA, in 1983, 1985, 1989, and 1986, respectively He is the John L. and Genevieve H. McCain Chair Professor in the PSU Department of Electrical Engineering. He is the Director of the Computational Electromagnetics and Antennas Research Lab (CEARL) http://cearl.ee.psu.edu/ as well as a member of the Communications and Space Sciences Lab (CSSL). He is also a faculty member of the Materials Research Institute (MRI) at PSU. He was a former Associate Editor of Radio Science and an Editor of the IEEE Antennas and Propagation Magazine. He holds eight patents, has authored more than 625 technical papers and proceedings articles, and is the author of 14 book chapters with several additional chapters currently in preparation. He edited a book entitled Frontiers in Electromagnetics (Piscataway, NJ, USA: IEEE Press, 2000). He has also contributed a chapter for a book entitled Electromagnetic Optimization by Genetic Algorithms (New York, NY, USA: Wiley Interscience, 1999) as well as for the book entitled Soft Computing in Communications (New York, NY, USA: Springer, 2004). He coauthored the book Genetic Algorithms in Electromagnetics (Hoboken, NJ, USA: Wiley/IEEE, 2007). He has also contributed an invited chapter on “Fractal Antennas” for the Antenna Engineering Handbook (New York, NY, USA: McGraw-Hill, 2007) as well as a chapter on “Ultra-Wideband Antenna Arrays” for a book entitled Frontiers in Antennas: Next Generation Design and Engineering (New York, NY, USA: McGraw-Hill, 2011). He has also contributed chapters for Numerical Methods for Metamaterial Design (New York, NY, USA: Springer, 2013), and Computational Electromagnetics (New York, NY, USA: Springer, 2014). His research interests include computational electromagnetic, antenna theory and design, phased arrays (including ultra-wideband arrays), microwave devices, wireless and personal communication systems (including on-body networks), wearable and e-textile antennas, RFID tag antennas, conformal antennas, reconfigurable antennas, frequency selective surfaces, electromagnetic wave interactions with complex media, metamaterials, electromagnetic bandgap materials, zero and negative index materials, transformation optics, nanoscale electromagnetics (including nanoantennas), fractal and knot electrodynamics, and nature-inspired optimization techniques (genetic algorithms, clonal selection algorithms, particle swarm, wind driven optimization, and various other evolutionary programming schemes). Dr. Werner is a member of the American Geophysical Union (AGU), URSI Commissions B and G, the Applied Computational Electromagnetics Society (ACES), Eta Kappa Nu, Tau Beta Pi and Sigma Xi. He is a Fellow of the IET (formerly IEE) and the ACES. He was presented with the 1993 Applied Computational Electromagnetics Society (ACES) Best Paper Award and was also the recipient of a 1993 International Union of Radio Science (URSI) Young Scientist Award. In 1994, he received the PSU Applied Research Laboratory Outstanding Publication Award. He was the recipient of a College of Engineering PSES Outstanding Research Award and Outstanding Teaching Award in March 2000 and March 2002, respectively. He was also presented with an IEEE Central Pennsylvania Section Millennium Medal. In March 2009, he received the PSES Premier Research Award. He was a coauthor of a paper published in the IEEE TRANSACTIONS ON ANTENNAS AND PROPAGATION, which received the 2006 R. W. P. King Award. He received the inaugural IEEE Antennas and Propagation Society Edward E. Altshuler Prize Paper Award and the Harold A. Wheeler Applications Prize Paper Award in 2011 and 2014, respectively. In 2015, he received the ACES Technical Achievement Award. He has also received several Letters of Commendation from the PSU Department of Electrical Engineering for outstanding teaching and research.